1. Field of the Invention
The present invention relates to the field of oscillators and modulation schemes therefor. More particularly, the present invention relates to the field of improved frequency control systems for oscillators with low phase noise.
2. Background and the Prior Art
Designers of digital radio frequency telecommunications systems such as the cellular data transmission networks now being contemplated and designed are limited by the Federal Communication Commission in the bandwidth and amplitudes they are allowed to use in various radio frequency bands to communicate data. Designers have an ever present need to transmit as much data as possible within the limitations of the bands given to them by the FCC. As a result, modulation schemes evolved from amplitude modulation where different amplitudes are used to represent information symbols (or digital states), to frequency modulation where different frequencies are used to represent information symbols (or digital states). From there, modulation schemes evolved from phase modulation where a particular phase is used to represent the information to QAM modulation schemes. The QAM modulation scheme uses the phase and amplitude of a carrier frequency relative to a stable frequency, single amplitude reference source to encode digital data. Because regulations imposed by the Federal Communication Commission limit bandwidth and amplitude that may be transmitted, QAM modulation schemes, and many coherent phase shift keying and frequency shift keying modulation schemes arose from the need for more efficient modulation schemes. FIG. 1 shows a symbolic diagram of a QAM modulation scheme.
In FIG. 1, axes 10 and 12 divide space into four quadrants. Spread throughout these four quadrants is a "constellation" of points or symbols, each of which represents a particular member of the encoded information to be transmitted. Each character, such as encoded ASCII A symbol, is encoded into symbol space by setting the phase angle of the carrier relative to the phase angle of the reference at a value equal to the angle between the vector connecting the origin to that point and the axis 10 and by setting the amplitude equal to the amplitude of said vector. Thus, the vector 14 (a phasor vector) may represent an encoded ASCII A symbol while vector 16 represents an encoded ASCII B symbol. An interesting thing about QAM and many other modulation schemes is that the more points that are in the symbol constellation, the easier it is to generate error correction and detection (ECC) bits that detect and correct errors when they occur. Note however, that the greater the number of points in the constellation, the smaller is the angle which exists between adjacent points in the constellation. This is also true for other modulation schemes such as FSK and PSK. This can cause occasional errors in discriminating between different points or information symbols in the constellation if the phase noise of either the transmitter or receiver causes a phase error which is too large. Typical constellations on modern QAM modulation schemes consist of 64 points. Such schemes can only tolerate phase errors of four degrees before data errors occur. There is a trend toward constellations having even more points such as 128 points or as many as 264 points to increase channel efficiency still further. These modulation schemes are demodulated by generation of a coherent carrier in the receiver that is synchronized with the oscillator in the transmitter.
The concept of phase noise is illustrated in FIG. 2. There the statistical distribution of values of amplitude and frequency or instantaneous phase of the carrier frequencies emitted by a local oscillator, which is typically a voltage controlled oscillator, is shown by curve 20. Curve 22 in FIG. 2 represents the statistical distribution of amplitude and frequency or instantaneous phase of the reference frequencies emitted by a crystal controlled reference source. Ideally, both these curves would be a single vertical line at the desired frequency were it not for the existence of phase noise. The vertical amplitude axis is proportional to the probability of carrier energy and the horizontal axis is frequency or instantaneous phase where frequency is simply equal to the rate of change of phase. Phase noise means that the phase of a sinusoidal signal relative to some fixed phase reference represented by line 25 is not fixed and jumps around with some statistical distribution. Line 25 in FIG. 2 represents a pure sinusoidal tone having an amplitude A.sub.1 and a fixed phase and frequency F.sub.ref to which the local oscillator frequency is to be synchronized coherently in phase. The flat parts of curve 22 represent phase noise which results from the fact that the active devices in a crystal controlled reference oscillator are not perfect and cause some modulation of the pure sinusoid in the form of shifts in the frequency away from the frequency of the pure sinusoidal tone. These shifts occur with the larger shifts occurring with decreasing statistical probability. In other words, at any particular instant in time, the output frequency of the crystal reference source will have the amplitude of the signal represented by line 25 but will have some frequency which is offset from F.sub.ref by some delta F with decreasing probability as the shift becomes larger. In other words, point 26 on curve 22 means that the probability or percentage of time that the actual output frequency of the crystal reference source has amplitude A.sub.1 and frequency F.sub.2 is greater than the probability represented by point 24 that the actual output frequency of the crystal reference source will have amplitude A.sub.1 and frequency F.sub.1. The Q of the oscillator also is a contributor to phase noise with higher Q oscillators causing narrower phase noise curves about the pure sinusoidal tone represented by line 25.
Curve 20 represents a typical phase noise curve for a microwave oscillator such as a YIG oscillator or a dielectric resonant oscillator. Each species of oscillator has its own peculiar phase noise characteristic. Note that for small frequency delta, the phase noise of the microwave oscillator is higher than that of the multiplied crystal controlled reference source, meaning that the probability of the frequency output by the local oscillator being some frequency different than the frequency F.sub.ref decreases with increasing delta F but is greater than the corresponding probability of an output of the same frequency for the crystal reference source. Note that the situation reverses itself at delta F.sub.2 or a frequency of F.sub.1. The areas under the curves 20 and 22 are the relative or RMS phase noises of the local oscillator and the crystal controlled reference sources, respectively.
Typically, local oscillators in the prior art have their output frequencies and phase locked to the frequency and phase of a reference source by a phase locked loop. The bandwidth or frequency gap between frequencies F.sub.1 and F.sub.3 in FIG. 2 is significant. This is because the band of frequencies between F.sub.1 and F.sub.3 in FIG. 2 represents the band wherein it is desirable to lock the frequency and phase of the local oscillator output signal to the frequency and phase of the crystal controlled reference signal for minimum phase noise. Outside the band of frequencies between F.sub.1 and F.sub.3, the phase noise of the microwave oscillator is better than the phase noise of the crystal controlled reference signal, and it is not desirable to lock the frequency and phase of the local oscillator output signal to the frequency and phase of the crystal controlled reference source.
Because the phase of the encoded symbol relative to the phase of the reference source (carrier or local oscillator) contains the information being sent, phase noise is a large problem because phase noise represents a source of errors. For example, in a QAM modulation scheme with a 64 point constellation, a phase error of four degrees or greater can lead to an incorrect symbol decision or bit error. If the error burst exceeds the range of receiver correction ability, then the frame of data containing the error or error burst will have to be retransmitted resulting in waste of valuable resources. Therefore it is extremely important in radio frequency digital data transmission to control the frequency and phase of the local oscillator so as to minimize the area under the composite phase noise curve of the local oscillator. A composite phase noise curve is a curve like the solid line curve shown at 30 in FIG. 3A. This composite phase noise curve 30 results from the effect of a phase locked loop controlling the frequency and phase of the local oscillator to the phase noise curve of the crystal controlled reference. The normal phase noise curve of the local oscillator is shown as dashed line curve at 34, and the phase noise curve of the crystal controlled reference source is shown as a dotted line curve at 32. Note how the composite phase noise curve conforms to the phase noise curve of the crystal controlled reference source within the frequency band between frequencies F.sub.1 and F.sub.3, but is allowed to generally assume its own phase noise characteristic outside the frequency band between frequencies F.sub.1 and F.sub.3.
Several phenomenon which cause phase noise related errors have been noticed in the prior art. One problem is perturbuations in the phase noise curves at specific frequencies which are thought to result from mechanical resonance frequencies in the packaging of the local oscillator or the cavities containing the YIG or dielectric material. FIG. 4 represents a phase noise curve for a local oscillator which has been phase locked to a crystal controlled reference source but which has mechanical package resonance phase noise disturbances. The peaks in phase noise at 50 and 52 are thought to represent the results of excitation of package mechanical resonance frequencies. These peaks will be hereafter referred to as package peaks, and have been found experimentally to occur at a mechanical resonance frequency of approximately 7 Kilo Hertz which causes deviations of the output frequency of the VCO away from the desired reference frequency by up to 7 kHz. These types of phase noise disturbances are also sometimes referred to as microphonics although microphonics is a broad term and also refers to any change or modulation in frequency caused by vibration. Microphonics cause the phase noise curve of an oscillator such as the phase noise curve 20 in FIG. 2 to broaden out and are a source of error of great concern in high efficiency modulation schemes. Much of the work in the prior art in stabilizing DRO and YIG oscillators using phase locked loops centers around the desire to eliminate phase noise caused by microphonics and package peaks.
Another type of phase noise disturbance that occurs are so called phase hits. These are a result of sudden changes in the phase of the local oscillator frequency. The cause of phase hits is sometimes mechanical relief of stress or strain within the package of the oscillator or by sudden unintentional changes in the capacitance of capacitors affecting the frequency of the local oscillator. Phase hits are serious problems because they can cause momentary loss of phase lock altogether which can result in error bursts which exceed the capacity of the ECC bits to detect and correct.
Both of these types of phase noise disturbances must be eliminated in a high quality oscillator sources for digital data transmission by high efficiency radios.
3. The Problem with the Prior Art Approaches
Digital telecommunication systems must have error rates down in the low parts per million range to be considered reliable. The phase noise of an oscillator source is the most important specification for designers of digital telecommunication systems. In the prior art, workers skilled in the art have used second order phase locked loops to attempt to reduce error bursts from package peaks, microphonics and phase hits. The assignee has found that second order PLL's will not adequately compensate for the errors caused by microphonics, phase hits and package peaks even though the manifestations of these phenomena on the phase noise curve is well within the bandwidth of the second order PLL. For example, two package peaks in the phase noise characteristic of a particular phase locked test oscillator were found at 7 kHz resulting from a package resonance excited by tapping on the table on which the oscillator sat. Even though the second order PLL bandwidth was 100 kHz, the package peaks were not eliminated from the phase noise curve. The prior art workers thought that their degraded phase noise characteristics were caused by some phase noise disturbances out very far from the center frequency so they kept increasing their loop bandwidth. However, the real problem was that the type of loop they were building was not adequate to eliminate, i.e., track out, the package peaks or microphonic effects even though the frequencies thereof were well within the bandwidth of the loop. The problem with the prior art second order loops was that they were not able to track and eliminate .omega./[(s+a).sup.2 +.omega..sup.2 ] terms in the Laplace domain which is what the package resonances and microphonics create. Some workers in the prior art have used third order loops to track and eliminate the effects of doppler shifts in frequency which have 1/s.sup.2 Laplace domain expressions. However, as will be discussed more fully below, these third order loops are inadequate to eliminate the microphonics problems though they have a third order polynomial denominator in their open loop transfer function they are inappropriate for microphonic disturbances.
A second order loop means that the polynomial representing the overall transfer function of the PLL has no higher term than s.sup.2 in denominator of the Laplace transform notation. A typical phase lock loop stabilizing a VCO local oscillator is shown in FIG. 5. A crystal controlled reference 60 generates the low phase noise reference signal on line 62 typically having the phase noise characteristics shown in FIG. 3A at curve 22. A nonlinear phase detector 64 receives the reference signal on line 62 and a feedback signal on line 66 which has the frequency and phase noise characteristics of the VCO local oscillator 68. The VCO typically operates at microwave frequecies such as 6 GHz, while the reference frequency is typically down around 100 Mhz.
The phase detector 64 generates a phase error signal on line 68 which is filtered by a loop filter 70 which typically takes the form of a single integrator such as an operational amplifier having a capacitor in its feedback network so as to integrate the incoming signal. The integrated phase error signal is fed on line 72 into the frequency control input of the VCO 68. Typically a second order loop will have a single integrator as the loop filter 70 which results in a second order transfer function because there is also an inherent integrator in the VCO. It takes at least a second order loop to track and correct for variations in both frequency and phase.
The loop filter 70 has a frequency response curve which in part defines a filter bandwidth. If the open loop gain is established such that the loop has large gain and does not roll off to the unity gain points until the intersection frequencies F.sub.1 and F.sub.3 are reached, then the closed loop response will track and eliminate any phase or frequency errors within the range of frequencies between F.sub.1 and F.sub.3, and the composite phase noise curve for the PLL shown in FIG. 5 will have the approximate shape of curve 30 in FIG. 3A. The area of curve 30 between frequencies F.sub.1 and F.sub.5 and the area between frequencies F.sub.3 and F.sub.4 represent areas where the loop gain is low. The closed loop transfer function of the PLL of FIG. 5 is essentially 1 when the open loop gain is large so any signal that appears at the output will be locked in frequency and phase to the frequency which appears at the input, i.e., the crystal controlled reference signal. Since the open loop gain is large inside the bandwidth of the loop, the loop will stay locked to the crystal reference frequency and phase inside the loop bandwidth. Outside the loop bandwidth, the open loop gain falls below the unity gain point, and the output signal will have the frequency and phase characteristics of the free running VCO, and that is why curve 30 in FIG. 3A looks like it does. The VCO will run at the crystal controlled reference frequency, but there will be no phase noise reduction outside the loop bandwidth.
Workers in the art have generally tried to use second order loops instead of third order loops to stabilize VCO frequency and phase characteristics because third order loops were thought to be unstable and difficult or impossible to get to converge and lock on the reference phase and frequency. U.S. Pat. No. 3,740,671 to Crow et al. teaches this conventional wisdom at Col. 2, lines 21-23, but teaches use of a third order loop which overcomes these limitations. The teachings of this patent are hereby incorporated by reference. In general, third order loops have been used in the prior art to track carrier signals which were shifting in frequency because of doppler effects. The following U.S. patents teach such applications of third order loops, frequently in the satellite tracking receiver context: 5,034,748; 4,860,321; 4,706,263; 3,878,522; 3,740,671, the teachings of which are hereby incorporated by reference. Third order loops have also appeared in such applications as a spread spectrum communication system (U.S. Pat. No. 4,841,544), a fast settling phase lock loop (U.S. Pat. No. 4,937,536) a beam riding missile guidance system (U.S. Pat. No. 4,516,743) and a narrowband phase modulation system (U.S. Pat. No. 4,053,834). The teachings of all these patents are hereby incorporated by reference.
In a chapter called "Automatic Phase Control" from the book PRINCIPLES OF COHERENT COMMUNICATIONS by Viterbi (McGraw Hill, New York), the author teaches the nonlinear sinusoidal analysis of the phase lock loop acquisition and lock process. Viterbi teaches that PLLs will only lock when a VCO output frquency and phase error result in a point in the phase plane plot of convergence trajectories that lie on a trajectory that are within a PLL "pull-in" range. Viterbi also taught that third order PLLs had rather lackluster performance characteristics under the conditions he taught.
The Crow et al. patent mentioned above incorporates the teachings of a paper published by Tausworthe and Crow of the Jet Propulsion Laboratory in Pasadena, Calif. entitled, "Practical Design of Third-Order Phase-Locked Loops". That paper teaches hybrid PLL's which acquire lock as second-order loops and then convert to third order loop operation by addition of another pole to the loop transfer function to remove loop stress. Tausworthe and Crow taught that Viterbi was wrong in that if the convergence problems of third order loops were separated out and dealt with as a separate problem and the loop transfer function was optimized in a particular manner that third order loops could outperform second order loops. However, Tausworthe and Crow taught that in order to make a third order loop converge, the loop had to be started as a second order loop and then an extra integrator was added to convert the transfer function to third order operations after convergence. Tausworthe and Crow taught a transfer function which had a -18 dB/octave frequency response over a small band of frequencies and achieved better results than Viterbi. What they did not realize was what the applicant has discovered, i.e., that they could have achieved far better performance by extending the band of frequencies over which the -18 dB/octave frequency response existed.
FIG. 6 shows a typical microwave transmitter using the output of the stabilized and phase locked VCO signal. The encoded digital data to be transmitted enters on line 80 and gets mixed and QAM modulated in modulator 82 with a microwave carrier on line 84 from the output of the local oscillator VCO 68. The components of the PLL are the same as shown in FIG. 5 except that a multiplier 86 is shown to multiply the frequency of the crystal controlled reference 60 up into the microwave frequencies of the desired carrier frequency. Note that the multiplication factor also raises the level of the asymptotic portions 21 and 23 in FIG. 2 of the skirt of the phase noise characteristic of the crystal controlled reference source by an amount proportional to the logarithm of the multiplication factor. The output of the modulator 82 is then applied to a microwave antenna 90. The VCO oscillator can be either a DRO or a YIG based oscillator or some other kind of oscillator.
It is the area underneath the composite phase noise curve which is of greatest significance to radio designers and operators of digital radio transmission systems since the integrating the phase noise curve gives this area in units of degrees of RMS phase noise jitter. Typical QAM modulation schemes modulate data out in the region to the right of frequency F.sub.4 in FIG. 3A. Therefore, it is very important to have a very small error under the composite phase noise curve in this region.
The effect of microphonic modulation including package resonance peaks and phase hit modulations are seen in FIG. 3A by inspection of curve 37 shown in dashed lines. These microphonic vibrations and phase hits can substantially broaden the phase noise curve and cause errors or bursts of error which exceed the range of detection and/or correction of the error correction bits.
FIG. 3B illustrates how these errors can occur. Suppose that the phase of the unmodulated carrier is .phi..sub.REF and the phase of one encoded symbol in a QAM scheme is .phi..sub.REF +5.degree. while the phase of an adjacent symbol in a QAM scheme is .phi..sub.REF -5.degree.. Suppose also that the decoding scheme involves two filters, one for each of these codes. One of these filters monitors for energy in phase window W.sub.1, while the other filter monitors for energy in phase window W.sub.2. Assume that the information encoded by a segment of carrier modulated in phase window W.sub.1 is an ASCII A, while the information encoded by a segment of carrier modulated in phase window W.sub.2 is an ASCII B. Now suppose an ASCII A is intended to be transmitted, but because of phase noise, in the particular time window in which the receiver is looking for what was supposed to be an ASCII A, the phase noise causes the carrier to actually have a phase somewhere in the crosshatched region shown at 90. In this example, the filter monitoring phase window W.sub.2 will pick up energy while the filter monitoring phase window W.sub.1 will not pick up any energy. The result will be an erroneous decoding of an ASCII B when an ASCII A was supposed to have been decoded.
In an attempt to reduce the effects on phase noise of phase hits, package peaks and microphonics, workers in the prior art have generally tried to make a tradeoff by increasing the loop bandwidth of second order PLL's to attempt to control the effects of microphonics at the expense of letting in more phase noise originating with the active devices. The difficulty with this approach is that while it reduces the effect of the microphonics and other sources of phase noise, it makes the overall phase noise worse. Further, it does not eliminate the errors caused by phase hits and microphonics or package peaks totally. The reason overall phase noise is made worse is because when the bandwidth is increased to encompass frequencies outside the range of between F.sub.1 and F.sub.3, the undesirable phase noise characteristics of the crystal controlled reference source outside F.sub.1 and F.sub.3 are incorporated into the final composite phase noise curve. If the phase noise could be controlled effectively without increasing the bandwidth of the PLL to include frequencies outside the range of frequencies between F.sub.1 and F.sub.3, then the more desirable phase noise characteristics of the VCO itself would control the composite phase noise characteristic curve in this range. Such a desired composite phase noise curve is shown at 30 in FIG. 3A.
In today's crowded cities, cellular phone systems are reaching saturation in channel usage in some areas. Because only a limited bandwidth is made available by the FCC for cellular traffic, the signals from cellular devices are split among a finite number of channels. Workers in this art have found that the cellular transmitters on building tops and hilltops need to be spread apart by a certain distance and a particular distance needs to be maintained between transmitters transmitting on the same channel frequency to avoid interference between the transmitters. This limits the number of channels available and limits the number of cellular devices that can be in use in any particular area at any particular time. With the increasing popularity of cellular modems, personal communicators, pagers and other purely digital devices, the limitation on available channel capacity poses a problem for further expansion of the cellular system. Since cellular telephones have proven themselves to be about 12 times more popular than original estimates, a serious shortfall in cellular capacity is beginning to develop.
The invention is applicable to both DRO and YIG oscillators as well as other types of oscillators. The invention also makes possible other new types of modulation schemes such as new types of spread spectrum communications or codivision multiplexing. Such modulation schemes have low probability of interception or detection which is useful in certain applications. The invention makes possible a greater degree of frequency re-use within a band because the frequency of the carrier never has to stop changing when the invention is in use because the third order PLL of the invention can track a carrier frequency which is continuously moving. In prior art spread spectrum communication modulation schemes, the frequency or the phase of the carrier stops changing from time to time such that the modulation scheme involves a series of hops from one frequency or phase to another. Thus, if two transmitters within range of each other both simultaneously hop to the same frequency and dwell there for some short time, there will be an error burst or interference resulting from the signals interferring with each other. With the invention, the frequency or phase dwell time at any particular setting is reduced to zero so the degree of possible interference is reduced to the short time when two transmitters within range of each other sweep past each other in frequency or phase. This allows more signals to be transmitted through a given band of frequencies than was otherwise possible because the transmitters never stop on any particular frequency. This means that the probability is diminished that any two transmitters, which are close enough together physically to interfere, will be transmitting on any particular frequency for a long enough time to cause an error burst which is long enough to be uncorrectable by the ECC bits. The possibility of the availability of these new spread spectrum modulation schemes thus promises to ease the developing shortfall in cellular communication channel capacity. As a fringe benefit, cellular communications will become more secure so interception of private cellular communications will be less likely. This will make cellular communication even more desirable as a mode of communication. These spread spectrum communication schemes with continuously moving carrier frequency require PLL's with great capacity to track and lock onto the phase of a continuously moving reference source.
Accordingly a need has arisen for a PLL design which can track the crystal controlled reference source even in the presence of phase hits, microphonics or package resonances and eliminate these sources of phase noise errors. This will maximize the ability to communicate with the limited frequency resources.